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sa571 compandor product specification 1997 aug 14 integrated circuits ic17 data handbook
philips semiconductors product specification sa571 compandor 2 1997 aug 14 853-0812 18285 description the sa571 is a versatile low cost dual gain control circuit in which either channel may be used as a dynamic range compressor or expandor. each channel has a full-wave rectifier to detect the average value of the signal, a linerarized temperature-compensated variable gain cell, and an operational amplifier. the sa571 is well suited for use in cellular radio and radio communications systems, modems, telephone, and satellite broadcast/receive audio systems. features ? complete compressor and expandor in one ichip ? temperature compensated ? greater than 110db dynamic range ? operates down to 6vdc ? system levels adjustable with external components ? distortion may be trimmed out ? dynamic noise reduction systems ? voltage-controlled amplifier pin configuration rect cap 1 rect in 1 ag cell in 1 gnd rect cap 2 ag cell in 2 rect in 2 v cc d, and n packages 1 1 2 3 4 5 6 7 8 9 10 11 12 13 14 16 15 inv. in 1 res. r 3 1 output 1 thd trim 1 inv. in 2 res. r 3 2 output 2 thd trim 2 top view note: 1. sol - released in large so package only. sr00675 figure 1. pin configuration applications ? cellular radio ? high level limiter ? low level expandorenoise gate ? dynamic filters ? cd player ordering information description temperature range order code dwg # 16-pin plastic small outline large (sol) -40 to +85 c sa571d sot162-1 16-pin plastic dual in-line package (dip) -40 to +85 c sa571n sot38-4 block diagram variable gain d g in rect in v ref thd trim r2 20k r1 10k rect cap r3 r3 20k r4 30k 1.8v inverter in output + rectifier sr00676 figure 2. block diagram
philips semiconductors product specification sa571 compandor 1997 aug 14 3 absolute maximum ratings symbol parameter rating units v cc maximum operating voltage 571 18 vdc t a operating ambient temperature range sa -40 to +85 c p d power dissipation 400 mw ac electrical characteristics v cc = +6v, t a = 25 c; unless otherwise stated. limits symbol parameter test conditions sa571 5 units min typ max v cc supply voltage 6 18 v i cc supply current no signal 3.2 4.8 ma i out output current capability 20 ma sr output slew rate .5 v/ m s gain cell distortion 2 untrimmed trimmed 0.5 0.1 2.0 % resistor tolerance 5 15 % internal reference voltage 1.65 1.8 1.95 v output dc shift 3 untrimmed 30 150 mv expandor output noise no signal, 15hz-20khz 1 20 60 m v unity gain level 6 1khz -1.5 0 +1.5 dbm gain change 2, 4 0.1 db reference drift 4 +2, -25 +20, -50 mv resistor drift 4 +8, -0 % tracking error (measured relative to value at unity gain) equals [v o - v o (unity gain)] db - v 2 dbm rectifier input, v 2 = +6dbm, v 1 = 0db v 2 = -30dbm, v 1 = 0db +0.2 +0.2 -1, +1.5 db channel separation 60 db notes: 1. input to v 1 and v 2 grounded. 2. measured at 0dbm, 1khz. 3. expandor ac input change from no signal to 0dbm. 4. relative to value at t a = 25 c. 5. electrical characteristics for the sa571 only are specified over -40 to +85 c temperature range. 6. 0dbm = 775mv rms .
philips semiconductors product specification sa571 compandor 1997 aug 14 4 circuit description the sa571 compandor building blocks, as shown in the block diagram, are a full-wave rectifier, a variable gain cell, an operational amplifier and a bias system. the arrangement of these blocks in the ic result in a circuit which can perform well with few external components, yet can be adapted to many diverse applications. the full-wave rectifier rectifies the input current which flows from the rectifier input, to an internal summing node which is biased at v ref . the rectified current is averaged on an external filter capacitor tied to the c rect terminal, and the average value of the input current controls the gain of the variable gain cell. the gain will thus be proportional to the average value of the input signal for capacitively-coupled voltage inputs as shown in the following equation. note that for capacitively-coupled inputs there is no offset voltage capable of producing a gain error. the only error will come from the bias current of the rectifier (supplied internally) which is less than 0.1 m a. g |v in v ref |avg r 1 or g |v in |avg r 1 the speed with which gain changes to follow changes in input signal levels is determined by the rectifier filter capacitor. a small capacitor will yield rapid response but will not fully filter low frequency signals. any ripple on the gain control signal will modulate the signal passing through the variable gain cell. in an expander or compressor application, this would lead to third harmonic distortion, so there is a trade-off to be made between fast attack and decay times and distortion. for step changes in amplitude, the change in gain with time is shown by this equation. g(t) (g initial g final ) e t g final ; 10k x c rect the variable gain cell is a current-in, current-out device with the ratio i out /i in controlled by the rectifier. i in is the current which flows from the d g input to an internal summing node biased at v ref . the following equation applies for capacitively-coupled inputs. the output current, i out , is fed to the summing node of the op amp. i in v in v ref r 2 v in r 2 a compensation scheme built into the d g cell compensates for temperature and cancels out odd harmonic distortion. the only distortion which remains is even harmonics, and they exist only because of internal offset voltages. the thd trim terminal provides a means for nulling the internal offsets for low distortion operation. the operational amplifier (which is internally compensated) has the non-inverting input tied to v ref , and the inverting input connected to the d g cell output as well as brought out externally. a resistor, r 3 , is brought out from the summing node and allows compressor or expander gain to be determined only by internal components. the output stage is capable of 20ma output current. this allows a +13dbm (3.5v rms ) output into a 300 w load which, with a series resistor and proper transformer, can result in +13dbm with a 600 w output impedance. a bandgap reference provides the reference voltage for all summing nodes, a regulated supply voltage for the rectifier and d g cell, and a bias current for the d g cell. the low tempco of this type of reference provides very stable biasing over a wide temperature range. the typical performance characteristics illustration shows the basic input-output transfer curve for basic compressor or expander circuits. +20 +10 0 10 20 30 40 50 60 70 80 40 30 20 10 0 +10 compressor output level or expandor input level (dbm) compressor input level or expandor output level (dbm) sr00677 figure 3. basic input-output transfer curve typical test circuit 20k 10k 13 3.14 2.2 2.15 4 1.16 2.2 5.12 8.2k 8.9 200pf 30k 20k 7.10 6.11 v 1 v 2 v o v cc = 15v v ref d g 10 m f 0.1 m f 2.2 m f sr00678 figure 4. typical test circuit introduction much interest has been expressed in high performance electronic gain control circuits. for non-critical applications, an integrated circuit operational transconductance amplifier can be used, but when high-performance is required, one has to resort to complex discrete circuitry with many expensive, well-matched components.
philips semiconductors product specification sa571 compandor 1997 aug 14 5 this paper describes an inexpensive integrated circuit, the sa571 compandor, which offers a pair of high performance gain control circuits featuring low distortion (<0.1%), high signal-to-noise ratio (90db), and wide dynamic range (110db). circuit background the sa571 compandor was originally designed to satisfy the requirements of the telephone system. when several telephone channels are multiplexed onto a common line, the resulting signal-to-noise ratio is poor and companding is used to allow a wider dynamic range to be passed through the channel. figure 5 graphically shows what a compandor can do for the signal-to-noise ratio of a restricted dynamic range channel. the input level range of +20 to -80db is shown undergoing a 2-to-1 compression where a 2db input level change is compressed into a 1db output level change by the compressor. the original 100db of dynamic range is thus compressed to a 50db range for transmission through a restricted dynamic range channel. a complementary expansion on the receiving end restores the original signal levels and reduces the channel noise by as much as 45db. the significant circuits in a compressor or expander are the rectifier and the gain control element. the phone system requires a simple full-wave averaging rectifier with good accuracy, since the rectifier accuracy determines the (input) output level tracking accuracy. the gain cell determines the distortion and noise characteristics, and the phone system specifications here are very loose. these specs could have been met with a simple operational transconductance multiplier, or ota, but the gain of an ota is proportional to temperature and this is very undesirable. therefore, a linearized transconductance multiplier was designed which is insensitive to temperature and offers low noise and low distortion performance. these features make the circuit useful in audio and data systems as well as in telecommunications systems. basic circuit hook-up and operation figure 6 shows the block diagram of one half of the chip, (there are two identical channels on the ic). the full-wave averaging rectifier provides a gain control current, i g , for the variable gain ( d g) cell. the output of the d g cell is a current which is fed to the summing node of the operational amplifier. resistors are provided to establish circuit gain and set the output dc bias. input level compression expansion output level noise +20 0db 40 80 20 0db 40 80 sr00679 figure 5. restricted dynamic range channel the circuit is intended for use in single power supply systems, so the internal summing nodes must be biased at some voltage above ground. an internal band gap voltage reference provides a very stable, low noise 1.8v reference denoted v ref . the non-inverting input of the op amp is tied to v ref , and the summing nodes of the rectifier and d g cell (located at the right of r 1 and r 2 ) have the same potential. the thd trim pin is also at the v ref potential. figure 7 shows how the circuit is hooked up to realize an expandor. the input signal, v in , is applied to the inputs of both the rectifier and the d g cell. when the input signal drops by 6db, the gain control current will drop by a factor of 2, and so the gain will drop 6db. the output level at v out will thus drop 12db, giving us the desired 2-to-1 expansion. figure 8 shows the hook-up for a compressor. this is essentially an expandor placed in the feedback loop of the op amp. the d g cell is setup to provide ac feedback only, so a separate dc feedback loop is provided by the two r dc and c dc . the values of r dc will determine the dc bias at the output of the op amp. the output will bias to: v out dc 1 r dc1 r dc2 r 4 v cc pin 13 gnd pin 4 output 7,10 v ref 1.8v r 4 30k 1,16 c rect r 1 10k 2,15 rect in g in 3,14 20k r 2 20k r 3 6,11 5,12 inv in r 3 thd trim 8,9 ig d g sr00680 figure 6. chip block diagram (1 of 2 channels) v in v out v ref d g *c in1 *c in2 *c rect + r 3 r 4 r 1 r 2 gain 2r 3 v in (avg) r 1 r 2 i b note: i b = 140 m a *external components sr00681 figure 7. basic expander v ref 1 r dctot 30k 1.8v the output of the expander will bias up to: v out dc 1 r 3 r 4 v ref v ref 1 20k 30k 1.8v 3.0v the output will bias to 3.0v when the internal resistors are used. external resistors may be placed in series with r 3 , (which will affect the gain), or in parallel with r 4 to raise the dc bias to any desired value.
philips semiconductors product specification sa571 compandor 1997 aug 14 6 notes: gain r 1 r 2 i b 2r 3 v inavg 1 2 i b = 140 m a external components v in c in c f r 1 r 2 r 3 v out d g * c rect * r dc * r dc * c dc * v ref r 4 sr00682 figure 8. basic compressor 10k c r i g r s r 1 v in v+ i = v in / r 1 sr00684 figure 9. rectifier concept circuit detailserectifier figure 9 shows the concept behind the full-wave averaging rectifier. the input current to the summing node of the op amp, v in r 1 , is supplied by the output of the op amp. if we can mirror the op amp output current into a unipolar current, we will have an ideal rectifier. the output current is averaged by r 5 , cr, which set the averaging time constant, and then mirrored with a gain of 2 to become i g , the gain control current. figure 10 shows the rectifier circuit in more detail. the op amp is a one-stage op amp, biased so that only one output device is on at a time. the non-inverting input, (the base of q 1 ), which is shown grounded, is actually tied to the internal 1.8v v ref . the inverting input is tied to the op amp output, (the emitters of q 5 and q 6 ), and the input summing resistor r 1 . the single diode between the bases of q 5 and q 6 assures that only one device is on at a time. to detect the output current of the op amp, we simply use the collector currents of the output devices q 5 and q 6 . q 6 will conduct when the input swings positive and q 5 conducts when the input swings negative. the collector currents will be in error by the a of q 5 or q 6 on negative or positive signal swings, respectively. ics such as this have typical npn b s of 200 and pnp b s of 40. the a 's of 0.995 and 0.975 will produce errors of 0.5% on negative swings and 2.5% on positive swings. the 1.5% average of these errors yields a mere 0.13db gain error. at very low input signal levels the bias current of q 2 , (typically 50na), will become significant as it must be supplied by q 5 . another low level error can be caused by dc coupling into the rectifier. if an offset voltage exists between the v in input pin and the base of q 2 , an error current of v os /r 1 will be generated. a mere 1mv of offset will cause an input current of 100na which will produce twice the error of the input bias current. for highest accuracy, the rectifier should be coupled into capacitively. at high input levels the b of the pnp q 6 will begin to suffer, and there will be an increasing error until the circuit saturates. saturation can be avoided by limiting the current into the rectifier input to 250 m a. if necessary, an external resistor may be placed in series with r 1 to limit the current to this value. figure 11 shows the rectifier accuracy vs input level at a frequency of 1khz. v+ 10k 10k q 1 q 2 q 3 q 4 q 7 q 5 q 6 q 8 q 9 c r r s r 1 d 1 i 1 i 2 v in v i g 2 v in avg r1 note: sr00683 figure 10. simplified rectifier schematic at very high frequencies, the response of the rectifier will fall off. the roll-off will be more pronounced at lower input levels due to the increasing amount of gain required to switch between q 5 or q 6 conducting. the rectifier frequency response for input levels of 0dbm, -20dbm, and -40dbm is shown in figure 12. the response at all three levels is flat to well above the audio range. error gain db +1 0 1 40 20 0 rectifier input dbm sr00685 figure 11. rectifier accuracy
philips semiconductors product specification sa571 compandor 1997 aug 14 7 0 3 10k 1meg input = 0dbm 20dbm 40dbm frequency (hz) gain error (db) sr00686 figure 12. rectifier frequency response vs input level variable gain cell figure 13 is a diagram of the variable gain cell. this is a linearized two-quadrant transconductance multiplier. q 1 , q 2 and the op amp provide a predistorted drive signal for the gain control pair, q 3 and q 4 . the gain is controlled by i g and a current mirror provides the output current. the op amp maintains the base and collector of q 1 at ground potential (v ref ) by controlling the base of q 2 . the input current i in (=v in /r 2 ) is thus forced to flow through q 1 along with the current i 1 , so i c1 =i 1 +i in . since i 2 has been set at twice the value of i 1 , the current through q 2 is: i 2 -(i 1 +i in )=i 1 -i in =i c2 . the op amp has thus forced a linear current swing between q 1 and q 2 by providing the proper drive to the base of q 2 . this drive signal will be linear for small signals, but very non-linear for large signals, since it is compensating for the non-linearity of the differential pair, q 1 and q 2 , under large signal conditions. q 1 q 2 q 3 q 4 i out i g i 1 i in i g v in i 2 r 2 note: i 2 (= 2i 1 ) 280 m a i g i in v in r 2 20k i 1 140 m a v+ v sr00687 figure 13. simplified d g cell schematic the key to the circuit is that this same predistorted drive signal is applied to the gain control pair, q 3 and q 4 . when two differential pairs of transistors have the same signal applied, their collector current ratios will be identical regardless of the magnitude of the currents. this gives us: i c1 i c2 i c4 i c3 i 1 i in i 1 i in plus the relationships i g =i c3 +i c4 and i out =i c4 -i c3 will yield the multiplier transfer function, i out i g i 1 i in v in r 2 i g i 1 this equation is linear and temperature-insensitive, but it assumes ideal transistors. 4 3 2 1 .34 6 0 +6 4mv 3mv 2mv 1mv input level (dbm) % thd v os = 5mv sr00688 figure 14. d g cell distortion vs offset voltage if the transistors are not perfectly matched, a parabolic, non-linearity is generated, which results in second harmonic distortion. figure 14 gives an indication of the magnitude of the distortion caused by a given input level and offset voltage. the distortion is linearly proportional to the magnitude of the offset and the input level. saturation of the gain cell occurs at a +8dbm level. at a nominal operating level of 0dbm, a 1mv offset will yield 0.34% of second harmonic distortion. most circuits are somewhat better than this, which means our overall offsets are typically about mv. the distortion is not affected by the magnitude of the gain control current, and it does not increase as the gain is changed. this second harmonic distortion could be eliminated by making perfect transistors, but since that would be difficult, we have had to resort to other methods. a trim pin has been provided to allow trimming of the internal offsets to zero, which effectively eliminated second harmonic distortion. figure 15 shows the simple trim network required. figure 16 shows the noise performance of the d g cell. the maximum output level before clipping occurs in the gain cell is plotted along with the output noise in a 20khz bandwidth. note that the noise drops as the gain is reduced for the first 20db of gain reduction. at high gains, the signal to noise ratio is 90db, and the total dynamic range from maximum signal to minimum noise is 110db. 3.6v v cc r 20k 6.2k to thd trim 200pf sr00689 figure 15. thd trim network
philips semiconductors product specification sa571 compandor 1997 aug 14 8 vca gain (0db) +20 output (dbm) 0 20 40 60 80 100 40 20 0 maximum signal level noise in 20khz bw 90db 110db sr00690 figure 16. dynamic range control signal feedthrough is generated in the gain cell by imperfect device matching and mismatches in the current sources, i 1 and i 2 . when no input signal is present, changing i g will cause a small output signal. the distortion trim is effective in nulling out any control signal feedthrough, but in general, the null for minimum feedthrough will be different than the null in distortion. the control signal feedthrough can be trimmed independently of distortion by tying a current source to the d g input pin. this effectively trims i 1 . figure 17 shows such a trim network. r-select for 3.6v 470k to pin 3 or 14 100k v cc sr00691 figure 17. control signal feedthrough operational amplifier the main op amp shown in the chip block diagram is equivalent to a 741 with a 1mhz bandwidth. figure 18 shows the basic circuit. split collectors are used in the input pair to reduce g m , so that a small compensation capacitor of just 10pf may be used. the output stage, although capable of output currents in excess of 20ma, is biased for a low quiescent current to conserve power. when driving heavy loads, this leads to a small amount of crossover distortion. q 1 q 2 q 4 q 3 i 1 i 2 q 6 d 1 d 2 q 2 c c +in in out sr00692 figure 18. operational amplifier resistors inspection of the gain equations in figures 7 and 8 will show that the basic compressor and expander circuit gains may be set entirely by resistor ratios and the internal voltage reference. thus, any form of resistors that match well would suffice for these simple hook-ups, and absolute accuracy and temperature coefficient would be of no importance. however, as one starts to modify the gain equation with external resistors, the internal resistor accuracy and tempco become very significant. figure 19 shows the effects of temperature on the diffused resistors which are normally used in integrated circuits, and the ion-implanted resistors which are used in this circuit. over the critical 0 c to +70 c temperature range, there is a 10-to-1 improve- ment in drift from a 5% change for the diffused resistors, to a 0.5% change for the implemented resistors. the implanted resistors have another advantage in that they can be made the size of the diffused resistors due to the higher resistivity. this saves a significant amount of chip area. ?????? ?????? 1% error normalized resistance temperature 1.15 1.10 1.05 1.00 .95 40 0 40 80 120 low tc implanted resistor band 1k w / 140 w / diffused resistor sr00693 figure 19. resistance vs temperature
philips semiconductors product specification sa571 compandor 1997 aug 14 9 so16: plastic small outline package; 16 leads; body width 7.5 mm sot162-1
philips semiconductors product specification sa571 compandor 1997 aug 14 10 dip16: plastic dual in-line package; 16 leads (300 mil) sot38-4
philips semiconductors product specification sa571 compandor 1997 aug 14 11 philips semiconductors and philips electronics north america corporation reserve the right to make changes, without notice, in the products, including circuits, standard cells, and/or software, described or contained herein in order to improve design and/or performanc e. philips semiconductors assumes no responsibility or liability for the use of any of these products, conveys no license or title under a ny patent, copyright, or mask work right to these products, and makes no representations or warranties that these products are free from patent, copy right, or mask work right infringement, unless otherwise specified. applications that are described herein for any of these products are for illustrative purposes only. philips semiconductors makes no representation or warranty that such applications will be suitable for the specified use without further testing or modification. life support applications philips semiconductors and philips electronics north america corporation products are not designed for use in life support appl iances, devices, or systems where malfunction of a philips semiconductors and philips electronics north america corporation product can reasonab ly be expected to result in a personal injury. philips semiconductors and philips electronics north america corporation customers using or sel ling philips semiconductors and philips electronics north america corporation products for use in such applications do so at their own risk and agree to fully indemnify philips semiconductors and philips electronics north america corporation for any damages resulting from such improper use or sale. this data sheet contains preliminary data, and supplementary data will be published at a later date. philips semiconductors reserves the right to make changes at any time without notice in order to improve design and supply the best possible product. philips semiconductors 811 east arques avenue p.o. box 3409 sunnyvale, california 940883409 telephone 800-234-7381 definitions data sheet identification product status definition objective specification preliminary specification product specification formative or in design preproduction product full production this data sheet contains the design target or goal specifications for product development. specifications may change in any manner without notice. this data sheet contains final specifications. philips semiconductors reserves the right to make changes at any time without notice, in order to improve design and supply the best possible product. ? copyright philips electronics north america corporation 1997 all rights reserved. printed in u.s.a.

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